Knock control apparatus for internal combustion engines

ABSTRACT

In an internal combustion engine knock control apparatus for detecting knocking phenomenon of an engine to control the timing of ignition of the engine, an output signal from a knock sensor is clamped at a value higher than a predetermined voltage value and half-wave rectified to produce a background signal. The background signal is compared with a knock signal to detect a knock intensity and the ignition timing of the engine is controlled in accordance with the detected knock intensity.

The present invention relates to knock control apparatus for internal combustion engines and more particularly to an internal combustion engine knock control apparatus capable of accurately detecting knock signals irrespective of the engine speed and properly controlling the occurrence of knocking.

The occurrence of knocking in an engine results in the occurrence of knocking sound so that the running performance of the engine is deteriorated and also the power output of the engine is decreased by the occurrence of counter torque or the engine is damaged due to overheating of the engine. It is known that the knocking has a close relation with the ignition timing and the maximum engine power can be produced by setting the ignition timing or the ignition advance angle just before the point of knocking in view of the characteristics of the engine. Thus, since retarding the ignition advance angle as a result of the prevention of knocking has the adverse effect of decreasing the engine power output, the ignition timing must be controlled to occur just before the point of knocking. Particularly, in the case of a turbocharger engine, the compression ratio is high and therefore the optimum ignition timing must be ensured so as to maintain the maximum efficiency. For this purpose, the occurrence of knocking in the engine must be detected accurately and the proper comparison between a knock signal and a background level must be ensured.

Generally, the output of a knock sensor increases with increase in the engine speed and hence the background level (BGL) increases correspondingly. Thus, at high engine speeds the occurrence of knocking can be detected accurately.

However, if all the outputs from the knock sensor are averaged to provide the desired background level, the resulting background level becomes so high as to cause a situation in which the detection of light knock is impossible. Thus, as proposed in Japanese Laid-Open Patent Application No. 57-59063, for example, an attempt has been made to detect a knock signal from the output of a knock sensor and mask the knock signal thereby preventing the knock signal from being reflected in the background level. However, there is a disadvantage that if all of the knock signals are masked, the background level is decreased so that signals other than those caused by the next knocking (i.e., knock signals) are also discriminated as knock signals and the ignition timing of the engine is retarded thereby greatly deteriorating the accuracy of knock detection.

It is the primary object of the present invention to provide a knock control apparatus which overcomes the foregoing deficiencies in the prior art and is capable of improving the accuracy of knock detection.

Thus, in accordance with the invention the accuracy of knock detection is improved by clamping a background voltage at a value higher than a predetermined voltage value and subjecting to half-wave rectification thereby producing a background level signal.

The present invention will become more apparent from the following description taken in conjunction with the accompanying drawings, in which:

FIG. 1 is a block diagram showing the overall construction of an embodiment of the invention;

FIGS. 2 to 8 are detailed circuit diagrams of the individual, component parts shown in FIG. 1;

FIGS. 9(A)-(N) show a timing chart useful for explaining the operation of the apparatus according to the invention;

FIG. 10 is an engine speed-advance angle characteristic diagram of the apparatus according to the invention.

The present invention will now be described with reference to the illustrated embodiment.

FIG. 1 shows the overall construction of an apparatus according to an embodiment of the present invention.

In the Figure, the knock control apparatus of the invention includes a knock sensor 100 for detecting a knock signal, a knock control unit 200 responsive to the knock signal applied from the knock sensor 100 to generate a control signal for controlling the ignition timing of an ignition coil 600, a pickup coil 400 for detecting the spark timing of the ignition coil 600, and a contactless ignition device 500 for igniting the ignition coil 600 in response to the outputs from the pickup coil 400 and the knock control unit 200 and supplying a feedback signal to the knock control unit 200.

The knock control unit 200 receives the detection signal from the knock sensor 100 and the output signal from the contactless ignition device 500 and controls the contactless ignition device 500 in response to the occurrence of knocking thereby advancing or retarding the ignition timing.

The knock control unit 200 includes an amplifier 201 including an ignition noise cutoff circuit 202 having a gate for cutting off ignition noise in synchronism with the spark timing, a band-pass filter or BPF 204 for knock signal band-pass purposes, a half-wave rectifier circuit 205 for suitably amplifying and half-wave rectifying the input signal from the BPF 204, a knock signal clamping circuit 206 responsive to a signal fed back from a background level detecting circuit 207 for effecting a clamping operation to prevent a high knock signal from entering the half-wave rectified signal from the half-wave rectifier circuit 205 and affecting the background level, the background level (BGL) detecting circuit 207 for producing an average value of the half-wave rectified signal generated from the half-wave rectifier circuit 205, a comparator 208 for comparing the output voltage of the BPF 204 and the output voltage of the BGL detecting circuit 207 to generate a retard signal proportional to the knocking, a masking circuit 210 for masking and generating the output of the comparator 208 at the desired timing, a knock signal voltage converting circuit 211 for integrating the output of the masking circuit 210 and generating a voltage value corresponding to the retard signal proportional to the knocking, a fail safe circuit 209 for detecting a fault in the knock sensor 100 and generating a signal to forcibly retard the ignition timing, a monostable circuit 212 responsive to the signal from the contactless ignition device 500 to generate a signal of a fixed pulse width in synchronism with the interruption of the ignition coil 600 (i.e., in synchronism with the base current to a power transistor 503), an F-V generator 213 responsive to the output pulse of the monostable circuit 212 to generate a voltage value proportional to the engine speed, a speed detecting circuit 214 responsive to the output from the F-V generator 213 to generate a signal corresponding to the engine speed, and a reference voltage generating circuit 203.

On the other hand, the contactless ignition device 500 includes an amplifier 501 for reshaping the waveform of an output signal from the pickup coil 400, a retard circuit 502 responsive to the output voltage of the knock control unit 200 to control the ignition timing, and the power transducer 503 for inducing a high voltage in the secondary winding of the ignition coil 600.

Next, the individual circuits of the knock control unit 200 will be described in detail.

FIG. 2 shows the detailed circuit constructions of the knock sensor 100, the amplifier 201, the ignition noise cutoff circuit 202 and the band-pass filter or BPF 204.

More specifically, the knock sensor 100 is a capacitor-type sensor employing a piezoelectric element and it is effectively a parallel circuit of a capacitor C and a constant current source.

A resistor R₁ is connected to the positive terminal of the knock sensor 100 and connected to the other end of the resistor R₁ are a capacitor C₁, resistors R₂ and R₃, the cathode of a Zener diode ZD₁ and the collector of a transistor T₁. The other end of the capacitor C₁, the other end of the resistor R₂, the anode of the Zener diode ZD₁ and the emitter of the transistor T₁ are grounded and the base of the transistor T₁ is connected to the mono-stable circuit (OSM) 212 through a resistor R₆. Also, the negative input terminal of an operational amplifier OP₁ is connected to the other end of the resistor R₃ through a capacitor C₂. The operational amplifier OP₁ is negatively fed back through a resistor R₄ and resistors R₇ and R₈ are connected to the output terminal of the operational amplifier OP₁. The other end of the resistor R₇ is grounded and capacitors C₄ and C₅ and a variable resistor 9 are connected to the other end of the resistor R₇.

The amplifier 201 are formed by the resistors R₁, R₂, R₃, R₄ and R₇, the capacitor C₂ and the operational amplifier OP₁ and the ignition noise cutoff circuit 202 is formed by the transistor T₁ and the resistor R₆.

The other end of the variable resistor R₉ is grounded and the negative input terminal of an operational amplifier OP₂ is connected to the other end of the capacitor C₄. The output terminal of the operational amplifier OP₂ is connected to the other end of the capacitor C₅. A negative feedback is applied to the operational amplifier OP₂ through a resistor R₁₀. Also, a capacitor C₃ is connected between the positive and negative input terminals of the operational amplifier OP₂. The positive input terminal of the operational amplifier OP₂ is connected to the positive input terminal of the operational amplifier OP₁, the positive input terminal of an operational amplifier OP₄ and a terminal S₆. Also, resistors R₁₁ and R₁₇ are connected to the output terminal of the operational amplifier OP₂. The other end of the resistor R₁₇ is grounded and the other end of the resistor R₁₁ is connected to a variable resistor R₁₂ and capacitors C₆ and C.sub. 7. The other end of the variable resistor R₁₂ is grounded and the other end of the capacitor C₆ is connected to the negative input terminal of the operational amplifier OP₄. A negative feedback is applied to the operational amplifier OP₄ through a resistor R₁₃. Also, the other end of the capacitor C₇ is connected to the output terminal of the operational amplifier OP₄. The positive input terminal of the operational amplifier OP₄ is connected to the positive input terminal of the operational amplifier OP₂ and the output terminal of the operational amplifier OP₄ is also connected to a resistor R₁₈ and a terminal S₅. The other end of the resistor R₁₈ is grounded.

The resistors R₈, R₉, R₁₀, R₁₁, R₁₂, R₁₃, R₁₇ and R₁₈, the capacitors C₃, C₄, C₅, C₆ and C₇ and the operational amplifiers OP₂ and OP₄ form the band-pass filter (BPF) 204. The BPF 204 is a two-stage filter.

FIG. 3 shows a detailed circuit diagram of the half-wave rectifier circuit 205 as well as the knock signal clamping circuit 206, the BGL detecting circuit 207 and the comparator 208 which constitute a feature of the present invention.

A capacitor C₈ is connected to the terminal S₅ and the other end of the capacitor C₈ is connected to the negative input terminal of an operational amplifier OP₆ through a resistor R₂₇. The negative input terminal of the operational amplifier OP₆ is also connected to a resistor R₂₅ and the anode of a diode D₁. The cathode of the diode D₁ is connected to the output terminal of the operational amplifier OP₆ and the anode of a diode D₂. The cathode of the diode D₂ is connected to a terminal S₇, the resistor R₂₅ and resistors R₂₆ and R₁₄. Also, the positive input terminal of the operational amplifier OP₆ is connected to the other end of the resistor R₂₆ and the terminal S₆.

The half-wave rectifier circuit 205 is formed by the resistors R₂₅, R₂₆ and R₂₇, the capacitor C₈, the diodes D₁ and D₂ and the operational amplifier OP₆.

The other end of the resistor R₁₄ is connected to the emitter of a PNP transistor T₂ and a resistor R₄₀. The collector of the PNP transistor T₂ is grounded and its base is connected to a resistor R₁₅ and the emitter of an NPN transistor T₃. The other end of the resistor R₁₅ is connected to the terminal S₆ and the collector of the NPN transistor T₃ is connected to a terminal S₁. The base of the NPN transistor T₃ is connected to the output terminal of an operational amplifier OP₁₀.

The resistors R₁₄ and R₁₅ and the transistors T₂ and T₃ form the knock signal clamping circuit 206.

On the other hand, the other end of the resistor R₄₀ is connected to the positive input terminal of the operational amplifier OP₁₀ and a capacitor C₁₆. The other end of the capacitor C₁₆ is connected to the terminal S₆. The negative input terminal of the operational amplifier OP₁₀ is connected to resistors R₄₃, R₄₄ and R₄₅. The other end of the resistor 43 is connected to a resistor R₄₁ and a variable resistor R₄₂. The other end of the resistor R₄₁ is connected to a terminal S₂. Also, the other end of the resistor R₄₄ is connected to the terminal S₆. The other end of the resistor R₄₅ is connected to the output terminal of the operational amplifier OP₁₀. The output terminal of the operational amplifier OP₁₀ is also connected to the base of the NPN transistor T₃ and the negative input terminal of a comparator CO₁.

The resistors R₄₀, R₄₁, R₄₂, R₄₃, R₄₄ and R₄₅, the capacitor 16 and the operational amplifier OP₁₀ form the background level (BGL) detecting circuit 207.

The positive input terminal of the comparator CO₁ is connected to the terminal S₅ and its output terminal is connected to a terminal R₁₆ and a terminal S₄. The other end of the resistor R₁₆ is connected to the terminal S₁.

The comparator CO₁ and the resistor R₁₆ form the comparator 208.

FIG. 4 shows the detailed circuit construction of the fail safe circuit 209 and the masking circuit 210.

In the Figure, resistors R₂₂ and R₂₄ and a terminal S₈ are connected to the terminal S₂ shown in FIG. 3. Also, the positive input terminal of an operational amplifier OP₇ is connected to the terminal S₇ shown in FIG. 3. The negative input terminal of the operational amplifier OP₇ is connected to resistors R₁₉ and R₂₀. The other end of the resistor 19 is connected to the terminal S₆ shown in FIG. 3 and a terminal S₉. The other end of the resistor R₂₀ is connected to the output terminal of the operational amplifier OP₇. The output terminal of the operational amplifier OP₇ is also connected through a resistor R₂₁ to the positive input terminal of a comparator CO₂ and the anode of a diode D₄.

The negative input terminal of the comparator CO₂ is connected to the other end of the resistor 22 and a resistor R₂₃. The other end of the resistor R₂₃ is connected to the terminal S₉. The output terminal of the comparator CO₂ is connected to the other end of the resistor R₂₄ and the anode of a diode D₇. The cathode of the diode D₇ is connected to a resistor R₈₉ whose other end is connected to a resistor R₃₁ and a capacitor C₉. The other end of the resistor R₃₁ is connected to a terminal S₁₂ and a resistor R₃₂.

The other end of the resistor R₃₂ and the other end of the capacitor C₉ are grounded.

The resistors R₁₉, R₂₀, R₂₁, R₂₂, R₂₃, R₂₄, R₃₁ and R₃₂, the diode C₇, the capacitor C₉, the operational amplifier OP₇ and the comparator CO₂ form the fail safe circuit 209.

On the other hand, the anode of a diode D₃ and the anode of a diode D₅ are connected to the terminal S₄ shown in FIG. 3. The cathode of the diode D₃ is connected to the collector of a transistor T₄ and the cathode of the diode D₄ is also connected to the collector of the transistor T₄. The emitter of the transistor T₄ is grounded and its base is connected to resistors R₂₈ and R₂₉. The other end of the resistor R₂₈ is connected to the terminal S₃ shown in FIG. 2 and the other end of the resistor R₂₉ is connected to a terminal S₁₁.

Also, the cathode of the diode D₅ is connected to a resistor R₃₀ and the base of a transistor T₅. The other end of the resistor R₃₀ and the emitter of the transistor T₅ are grounded and the collector of the transistor T₅ is connected to a terminal S₁₀.

The resistors R₂₈ and R₂₉, the diode D₃ and the transistor T₄ form the masking circuit 210.

FIG. 5 shows the detailed circuit construction of the monostable circuit (OSM) 212.

In the Figure, a resistor R₇₄ is connected to an input terminal IG for the ignition signal or the output signal from the power transistor 503 and the other end of the resistor R₇₄ is connected to a capacitor C₁₂, the cathode of a diode D₆ and the base of a transistor T₉. The other end of the capacitor C₁₂ and the anode of the diode D₆ are grounded. The emitter of the transistor T₉ is grounded through a resistor R₈₃ and its collector is connected to resistors R₇₅ and R₇₆. The other end of the resistor R₇₅ is connected to the terminal S₂ shown in FIG. 3 and the other end of the resistor R₇₆ is connected to the base of a transistor T₁₀ and a resistor R₈₁. The emitter of the transistor T₁₀ is grounded through a resistor R₈₃ and its collector is connected to the cathode of a diode D₈. The anode of the diode D₈ is connected to resistors R₇₈ and R₇₉. The other end of the resistor R₇₈ is connected to the terminal S₂ of FIG. 3 and the other end of the resistor R₇₉ is connected to the base of a transistor T₁₁ through a capacitor C₁₃. The transistor T₁₁ has its base connected to the terminal S₂ of FIG. 3 through a resistor R₈₀ and its emitter connected to the ground. The collector of the transistor T₁₁ is connected to the resistor R₈₁, a resistor R₈₂ and the terminal S₃ shown in FIG. 2. The other end of the resistor R₈₂ is connected to the terminal S₂ shown in FIG. 3.

The resistors R₇₄, R₇₅, R₇₆, R₇₈, R₇₉, R₈₀, R₈₁, R₈₂ and R₈₃, the capacitors C₁₂ and C₁₃, the diodes D₆ and D₈ and the transistors T₉, T₁₀ and T₁₁ form the monostable circuit 212.

FIG. 6 shows the detailed circuit constructions of the F-V generator 213 and the speed detecting circuit 214.

In the Figure, the base of a transistor T₆ is connected to the terminal S₃ shown in FIG. 5 through a resistor R₈₃. The transistor T₆ has its emitter grounded and its collector connected to a variable resistor R₈₄. The other end of the variable resistor R₈₄ is connected to the negative input terminal of an operational amplifier OP₁₄, a capacitor C₁₉ and a resistor R₈₅. The output terminal of the operational amplifier OP₁₄ is connected to the other end of the capacitor C₁₉ and the resistor R₈₅, respectively. The output terminal of the operational amplifier OP₁₄ is also connected to a resistor R₈₇, the negative input terminal of a comparator CO₃ and the negative input terminal of a comparator CO₅. The other end of the resistor R₈₇ is grounded. The positive input terminal of the operational amplifier OP₁₄ is connected to a terminal S₁₃.

The resistors R₈₃, R₈₅ and R₈₇, the variable resistor R₈₄, the capacitor C₁₉ and the operational amplifier OP₁₄ form the F-V generator 213.

On the other hand, the terminal S₂ shown in FIG. 3 is connected to resistors R₃₃, R₃₅, R₃₇, R₄₈, R₄₉, R₅₀ and R₅₁ and a terminal S₁₄. The other end of the resistor R₃₃ is connected to a resistor R₃₄, the positive input terminal of the comparator CO₃ and the anode of a diode D₉. The other end of the resistor R₃₄ is connected to the ground. Also, the cathode of the diode D₉ is connected to the output terminal of the comparator CO₃ through a resistor R₃₉. The output terminal of the comparator CO₃ is also connected to the other end of the resistor R₄₈ and a terminal S₁₅.

The other end of the resistor R₃₅ is connected to a resistor R₃₆, the positive input terminal of the comparator CO₄ and the anode of a diode D₁₀. The other end of the resistor R₃₆ is grounded. Also, the cathode of the diode D₁₀ is connected to the output terminal of the comparator CO₄ through a resistor R₄₆. The output terminal of the comparator CO₄ is also connected to the other end of the resistor R₄₉ and the terminal S₁₂ of FIG. 4 through a diode D₁₁.

The other end of the resistor R₃₇ is connected to a resistor R₃₈, the positive input terminal of the comparator CO₅ and a resistor R₄₇. The other end of the resistor R₃₈ is grounded and the other end of the resistor R₄₇ is connected to the collector of a transistor T₇. The transistor T₇ has its emitter grounded and its base connected to a terminal S₁₆ through a resistor R₈₈. Also, the output terminal of the comparator CO₅ is connected to the other end of the resistor R₅₀ and the base of a transistor T₁₂. The transistor T₁₂ has its emitter grounded and its collector connected to the other end of the resistor R₅₁ and the terminal S₁₆.

The speed detecting circuit 214 is formed by the resistors R₃₃, R₃₄, R₃₅, R₃₆, R₃₇, R₃₈, R₃₉, R₄₆, R₄₇, R₄₈, R₄₉, R₅₀, R₅₁ and R₈₈, the diodes D₉, D₁₀ and D₁₁, the transistors T₇ and T₁₂ and the comparators CO₃, CO₄ and CO₅.

FIG. 7 shows the detailed circuit construction of the knock signal voltage converting circuit 211.

In the Figure, connected to the terminal S₁₄ shown in FIG. 6 are resistors R₅₂, R₅₄, R₅₆, R₅₇, R₆₆, R₆₈ and resistor R₆₉ and a terminal S₈. The other end of the resistor R₅₂ is connected to a resistor R₅₃ and the collector of a transistor T₁₃. The base of the transistor T₁₃ is connected to the terminal S₁₂ shown in FIG. 4. The other end of the resistor R₅₃ is connected to the base of a transistor T₁₄. The emitter of the transistors T₁₃ and T₁₄ are grounded. Also, the collector of the transistor T₁₄ is connected to the other end of the resistor R₅₄, the cathode of a diode D₁₆ and a resistor R₅₅. The anode of the diode D₁₆ is connected through a reistor R₆₀ to the anode of a diode D₁₇ and the negative input terminal of an operational amplifier OP₁₁. The other end of the resistor R₅₅ is connected to the base of a transistor T₁₅. The transistor T₁₅ has its emitter grounded and its collector connected to the other end of the resistor R₅₆ and a resistor R₆₃. The other end of the resistor R₆₃ is connected to the base of a transistor T₈ and a resistor R₇₂. The other end of the resistor R₇₂ is connected to the terminal S₁₆. Also, the transistor T₈ has its emitter grounded and its collector connected to resistors R₈₉ and R₇₀. The other end of the resistor R₇₀ is grounded and the other end of the resistor R₈₉ is connected to the other end of the resistor R₆₉ and the negative input terminal of an operational amplifier OP₁₂.

On the other hand, the terminal S₁₅ shown in FIG. 6 is connected to the anode of a diode D₁₂ and a resistor R₆₂. The cathode of the diode D₁₂ is connected to the terminal S₁₁ shown in FIG. 4. Also, the other end of the resistor R₆₂ is connected through a diode D₁₃ to a resistor R₆₁, the cathode of a diode D₁₄ and the negative input terminal of the operational amplifier OP₁₁. The other end of the resistor R₆₁ is connected to the terminal S₁₀ shown in FIG. 4. Also, the anode of the diode D₁₄ is connected to the output terminal of the operational amplifier OP₁₂ through a resistor R₆₅. The positive input terminal of the operational amplifier OP₁₂ is connected to the output terminal of the operational amplifier OP₁₁, the positive input terminal of a comparator CO₆ and a resistor R₇₁. The other end of the resistor R₇₁ is grounded. Also, the negative input terminal of the comparator CO₆ is connected to the positive input terminal of the operational amplifier OP₁₁, the terminal S₁₃ shown in FIG. 6, the resistor R₆₈ and a resistor R₆₇. The other end of the resistor R₆₇ is grounded. The output terminal of the comparator CO₆ is connected to the other end of the resistor R₆₆ and a resistor R₆₄. The other end of the resistor R₆₄ is connected to the cathode of the diode D₁₇ whose anode is connected to the negative input terminal of the operational amplifier OP₁₁ and to the anode of the diode D₁₆ through the resistor R₆₀.

Also, the other end of the resistor R₅₇ is connected to resistors R₅₈ and R₅₉. The other end of the resistor R₅₈ is grounded. The other end of the resistor R₅₉ is connected to the negative input terminal of the operational amplifier OP₁₁. The negative input terminal and output terminal of the operational amplifier OP₁₁ are bridged by a parallel circuit of a capacitor C₁₈ and a diode D₁₅. Also, the output terminal of the operational amplifier OP₁₁ is connected to a resistor R₇₃ and the cathode of a Zener diode ZD₄. The other end of the resistor R₇₃ is connected to a capacitor C₁₁ and an output terminal SIG. Also, the anode of the Zener diode ZD₄ and the other end of the capacitor C₁₁ are grounded.

FIG. 8 shows the detailed circuit construction of the reference voltage generator 203.

In the Figure, a terminal V for the supply voltage (usually the battery voltage) is connected to a resistor R₈₄. The other end of the resistor R₈₄ is connected to a capacitor C₁₀, the cathode of a Zener diode ZD₂, a three-terminal regulator 50 (the Hitachi HA17M08) and the terminal S₁ shown in FIG. 3. The other end of the capacitor C₁₀, the anode of the Zener diode ZD₂ and the three-terminal regulator 50 are grounded. Also, the output terminal of the three-terminal regulator 50 is connected to a capacitor C₁₄, a resistor R₈₅ and the terminal S₈ shown in FIG. 4. The other end of the capacitor C₁₄ is grounded. The other end of the resistor R₈₅ is connected to a resistor R₈₆, a capacitor C₁₅ and the positive input terminal of an operational amplifier OP₁₅. The other end of the resistor R₈₆ and the other end of the capacitor C₁₅ are grounded. Also, the positive and negative input terminals of the operational amplifier OP₁₅ are connected through a capacitor C₁₇. The output terminal of the operational amplifier OP₁₅ is connected to a resistor R₈₇, a capacitor C₂₀, the negative input terminal of the operational amplifier OP₁₅ and the terminal S₉ shown in FIG. 4. The other end of the resistor R₈₇ and the capacitor C₂₀ are grounded.

The resistors R₈₅, R₈₆ and R₈₇, the capacitors C₁₄, C₁₇ and C₂₀ and the operational amplifier OP₁₅ form the reference voltage generator 203.

The operation of the knock control unit 200 will now be described.

Firstly, when a signal as shown in (A) of FIG. 9 is applied to the IG terminal shown in FIG. 5, in response to the high level of the signal the transistor T₉ is truned on and the transistor T₁₀ is turned off. When the transistor T₁₀ is turned off, a path including the power supply terminal S₁, the resistors R₇₈ and R₇₉, the capacitor C₁₃ and the base of the transistor T₁₁ is provided for the capacitor C₁₃. On the other hand, in response to the low level of the base signal the transistor T₉ is turned off and the transistor T₁₀ is turned on thereby providing a path including the power supply terminal S₁, the resistor R₈₀, the capacitor C₁₃, the resistor R₇₉, the diode D₈, the transistor T₁₀, the resistor R₈₃ and the ground. The two paths form a charging and discharging circuit for the capacitor C₁₃ and a pulse having a time width t₁ and synchronized with the spark timing as shown in (B) of FIG. 9 is generated at the collector of the transistor T₁₁. This signal is applied to the base of the transistor T₁ in the ignition noise cutoff circuit 202 to provide an ignition noise cutoff signal and the signal is also applied to the base of the transistor T₄ in the masking circuit 210 to perform the ignition noise cutoff function. Shown in (A) of FIG. 9 is the ignition timing waveform and this waveform signal is in fact the base signal to the power transistor 503 in the contactless ignition device 500 which will be described later. The power transistor 503 is turned on by the high level of the base signal and the power transistor 503 is turned off by the low level of the base signal. A spark is generated in the ignition coil in the course of switching between the on and off operations. The signal shown in (B) of FIG. 9 is the fixed-width pulse output signal of the monostable circuit 212 which receives the base signal so as to be triggered by its transition from the high to the low level thereby generating a pulse signal of the fixed width t₁. In other words, the signal (B) is the waveform at the collector of the transistor T₁₁.

Then, increasing the input impedance of the knock control apparatus tends to cause the superposition of disturbance noise. The typical of the disturbance noise is ignition noise (Ig noise) produced in synchronism with the ignition timing.

The ignition noise in the apparatus will now be described.

The base of the power transistor 503 is controlled by a pulse as shown in (A) of FIG. 9. The power transistor 503 is turned on when the pulse goes to the high level and is turned off when the pulse goes to the low level. In the course of switching between the turning on and off or when the transistor 503 is turned off, the secondary voltage in the ignition coil rises rapidly and primary noise is generated. Also, the rise in the secondary voltage causes breakdown in the insulation of the air layer of the plug and the ignition takes place. This ignition causes secondary noise. The secondary noise includes noise due to a capacitive discharge current flowing during the initial period of the ignition and noise due to an inductive discharge current flowing during the following period. The latter noise is a large noise source in the secondary noise. If the input impedance is increased, the primary noise and the secondary noise (the noise of the former) are superposed as disturbance noise on the output of the knock sensor adversely affecting the knock signal discrimination.

Such disturbance noise must be eliminated. This disturbance noise lasts for a period of about 50 to 60 μsec. Thus, it is necessary to mask the knock sensor output during this period. The ignition noise cutoff circuit 202 is provided for attaining this object. However, the actual masking period is preset to a time width which is sufficiently longer than the noise duration time, e.g., about 0.8 msec.

As a result, when a signal as shown in (C) of FIG. 9 is generated from the knock sensor 100, the amplitude of the signal is decreased as shown in (D) of FIG. 9 due to the resistance division by the resistors R₁ and R₂ and then supplied to the ignition noise cutoff circuit 202. The signal detected by the knock sensor 100 is one which varies between positive and negative with respect to a dc zero level as a reference. The ignition noise cutoff circuit 202 performs the ignition noise cutoff function mainly through the action of the transistor T₁. The transistor T₁ is turned on and off in response to the output of the monostable circuit 212. The monostable circuit 212 is triggered by the falling edge of the base signal of the power transistor 503 shown in (A) of FIG. 9 thereby generating a pulse of the masking time width. Shown in (B) of FIG. 9 is this output of the monostable circuit 202 and the time width t₁ represents the masking time width. The transistor T₁ is turned on only during the interval t₁ in which the output of the monostable circuit 212 goes to "1". Thus, during the interval t₁ the knock sensor output is short-circuited to the ground so that no input is applied to the operational amplifier OP₁ and the masking effect is produced for ignition noise masking purposes.

The ignition noise cutoff circuit 202 generates a signal as shown in (E) of FIG. 9.

The signal shown in (E) of FIG. 9 is amplified by the operational amplifier OP₁ and then subjected to the feedback action of the reference voltage generating circuit 203 thereby generating a signal as shown in (F) of FIG. 9 as a dc level signal (3.0 V) from the output terminal of the operational amplifier OP₁.

The amplification factor G of the operational amplifier OP₁ is given by ##EQU1##

The signal shown in (E) of FIG. 9 is applied to the band-pass filter (BPF) 204.

The BPF 204 emphasizes the knock signal (i.e., attenuates the other signals) and delivers it, that is, the circuit has a characteristic which slightly attenuates at higher frequencies than the knock signal due to knocking.

In the half-wave rectifier circuit 205, only the positive component is half-rectified by the action of the diodes D₁ and D₂ and applied to the knock signal clamping circuit 206. Then, after passing through the clamping circuit 206, the signal is integrated and smoothed by an integrating circuit formed by the resistor R₄₀ and the capacitor C₁₆ of the BGL detecting circuit 207, amplified by the operational amplifier OP₁₀ and then applied to the comparator 208.

The gain G1 of the half-wave rectifier circuit 205 is given by ##EQU2## and the gain G₂ of the amplifier formed by the resistors R₄₄ and R₄₅ and the operational amplifier OP₁₀ is given by ##EQU3##

Then, as regards the gain G3 of the integrator formed by the resistor R₄₀ and the capacitor C₁₆ responsive to the applied half-wave rectified signal, if E represents the half-wave peak voltage, then there holds ##EQU4## where t₀ ≦t≦t₁ : A=Esin (ωt)

t₁ ≦t≦t₂ : B=0

Therefore, the terminal voltage V_(c) (t) of the capacitor C₁₆ is given by ##EQU5## where t₀ ≦t≦t₁ ##EQU6## Thus, in the steady state we obtain

    V.sub.cl (t.sub.0)=V.sub.c2 (t.sub.2)

Therefore, V_(cl) (t₀) is given by ##EQU7## Substituting C₁₆ R₄₀ (=50 msec or over) and f (=5 kHz or over) in the equation (6), we obtain

    V.sub.cl (t.sub.0)=E·G.sub.3 ≃E/π(7)

In other words, we obtain the following with respect to the absolute value voltage (the terminal voltage of the capacitor C₁₆) V_(cl) (t₀)

    E=k·V.sub.cl (t.sub.0).

Assuming now that V_(B) respresents the signal voltage generated from the BPF 204, the voltage generated from the BGL detecting circuit 207 or the background voltage V_(BGL) is given by

    V.sub.BGL =V.sub.B ·1/k·G1·G2   (8)

The knock signal clamping circuit 206 constituting a feature of the invention will now be described.

Firstly, the signal generated as shown in (F) of FIG. 9 from the operational amplifier OP₁ of the amplifier 201 is applied to the BPF 204 so that the knock signal is emphasized and a signal as shown in (G) of FIG. 9 is generated from the operational amplifier OP₄ of the BPF 204. The signal generated from the BPF 204 is applied to the half-wave rectifier circuit 205. The half-wave rectifier circuit 205 applies to the knock signal clamping circuit 206 and the BGL detecting circuit 207 a signal which has been amplified by a given amount and half-wave rectified as shown in (H) of FIG. 9.

In the BGL detecting circuit 207, the applied signal is integrated and smoothed by the integrating circuit formed by the resistor R₄₀ and the capacitor C₁₆ and then amplified with an amplification factor G2 by the operational amplifier OP₁₀. The output signal of the operational amplifier OP₁₀ is a BGL signal. This BGL signal serves as an actuating signal for the knock signal clamping circuit 206.

In the knock signal clamping circuit 206, the knock signal is clamped at a BG voltage by the transistors T₂ and T₃. It has heretofore been known that if the whole knock signal (such signal exceeding a certain voltage value) were masked, the BG voltage would be decreased excessively thus discriminating a non-knocking signal as a knock as mentioned proviously. On the contrary, if a BG voltage is produced from the signal including a knock signal, the BG voltage would be increased excessively thus making it impossible to accurately detect the knock. To overcome these deficiencies in the prior art, the knock signal is clamped at the BG signal so that the BG signal assumes the proper value. In other words, the BG voltage is produced from a value at which the knock signal is clamped.

However, during the engine starting period the BG signal is the reference voltage (3 V) plus 0 V in the knock signal clamping circuit 206. As a result, if the knock signal is clamped at the BG voltage even during the starting period, the BG voltage indefinitely includes only the reference voltage and it is prevented from increasing any further. In other words, the knock control unit 200 itself is not actuated. Thus, the knock control unit 200 is operable at the actuating voltage of the knock signal clamping circuit 206 which is the reference voltage (3 V) plus 0.7 V. In other words, when the BG voltage is less than the reference voltage (3 V) plus 0.7 V, the knock signal clamping circuit 206 is not operated and thus the whole signal is used in the production of the BG voltage. Therefore, when the averaged value of the output signal from the half-wave rectifier circuit 205 exceeds the reference voltage (3 V) plus 0.7 V, the signal generated from the half-wave rectifier circuit 205 and having a value greater than the reference value (3 V) plus 0.7 V is clamped as shown in (I) of FIG. 9.

Thus, when the BG voltage exceeds the reference voltage (3 V) plus 0.7 V, the knock signal clamping circuit 206 comes into operation and a BG signal is produced by the clamped signal.

As a result, the BG signal generated from the BGL detecting circuit 207 and the signal generated from the BPF 204 are compared by the comparator 208 as shown in (J) of FIG. 9. The comparator CO₂ of the comparator 208 generates a rectangular waveform as shown in (K) of FIG. 9. This pulse signal is applied to the knock signal voltage converting circuit 211 through the masking circuit 210.

In the masking circuit 210, the transistor T₄ is turned on by the output signal from the monostable circuit 212 so that the current output of the comparator 208 flows to the ground through the transistor T₄ and it is masked. When the transistor T₄ is turned off, the output signal of the comparator 208 is applied to the transistor T₅ through the diode D₅ and the transistor T₅ is turned on.

In the knock signal voltage converting circuit 211 shown in FIG. 7, the operational amplifier OP₁₁, the capacitor C₁₈ and the diode D₁₅ form an output integrating circuit and the operational amplifier OP₁₂, the resistors R₆₅, R₆₉ and R₈₇ and the diode D₁₄ form a maximum voltage clamping circuit. Also, the comparator CO₅, the resistors R₆₇, R₆₄ and R₇₁ and the diode D₁₇ form a minimum voltage clamping circuit.

Now, in response to the output of the comparator 208 or the knock signal, the transistor T₅ is turned on in synchronism with the knock signal. Thus, as shown in (K) of FIG. 9, during the period of the knock signal pulse width t₀ (about 40 to 70 μsec) the transistor T₅ is turned on and a current i₁ flows from the operational amplifier OP₁₁ to the ground through the capacitor C₁₈, the resistor R₆₁ and the transistor T₅. At this time, the output voltage of the operational amplifier OP₁₁ is 3.0 V.

Thus, at this time the voltage rise rate per pulse (voltage rise/pulse) ΔV₁ of the operational amplifier OP₁₁ is obtained from ##EQU8## Here, the capacitance C represents the capacitance value of the capacitor C₁₈. As will be seen from the equation (10), the output voltage of the operational amplifier OP₁₁ increases in proportion to the number of knock pulses.

The Zener voltage of the Zener diode ZD₄ is 6 V. Also, the negative terminal of the operational amplifier OP₁₁ is at -3 V. As a result, each time a pulse is applied to the operational amplifier OP₁₁ from the comparator 208, the output voltage of the operational amplifier OP₁₁ decreases with the following voltage drop rate (drop voltage rate/period) ΔV₂ ##EQU9## Therefore ##EQU10## The voltage drop rate ΔV₂ is preset to about 1/50 of the voltage rise rate ΔV₁ in consideration of the power performances of the engine, such as, the engine torque and horsepower. The output of the output integrating circuit is clamped in such a manner that its maximum value is clamped at the clamping voltage of the maximum clamping circuit and its minimum value is clamped at the clamping voltage of the minimum clamping circuit.

The output integrating circuit is designed so that during the engine starting period the transistor T₄ is turned on by the output voltage of the 350 rpm detecting circuit of the speed detecting circuit 214 and the output of the comparator 208 is masked thus providing a specified spark advance characteristic (advance angles). In accordance with this spark advance characteristic, the output integrating circuit of the knock signal voltage converting circuit 211 generates a command and the retard circuit 502 effects the actual spark advance angle (retard angle) control. This retard circuit 502 may be of the same type as disclosed in U.S. patent application, Ser. No. 80,202, now U.S. Pat. No. 4,367,712, by Noboru Sugiura, filed Oct. 1, 1979 and assigned to the assignee of this application entitled "Ignition Timing Control System for Internal Combustion Engines".

The operation of the retard circuit 502 will now be described.

Generally, the ignition timing characteristic is of a relative nature and it is determined by the distributor and a certain operating mode determined by the ignition system used. Also, a maximum retard characteristic for knocking is predetermined so that the characteristic is used upon the occurrence of knocking. In FIG. 10 showing spark advance and retard characteristics, the solid line shows a minimum retard (minimum clamping voltage) characteristic in a certain operating mode and the broken line shows a maximum retard (maximum clamping voltage) characteristic under knocking conditions. At low engine speeds lower than 350 rpm, for example, the knock control is effected such that the maximum advance characteristic determined by the ignition timing characteristic is used. The reason for using this characteristic is to positively start the engine upon starting. In other words, if, during the engine start, the ignition timing is retarded, a counter torque is produced and the load on the starter is increased greatly. As a result, the driving current of the starter is increased abnormally and the engine is not operated thus causing a so-called starting failure. In order to prevent such starting failure, the maximum advance characteristic determined by the ignition timing characteristic is used during the starting operation of less than 350 rpm, for example.

FIG. 10 shows the characteristic of the retard circuit 502 which is required for accomplishing the above-mentioned features. As shown in the Figure, the circuit has a retard characteristic in the form of a fixed angle slope characteristic with respect to the output of the output integrating circuit in the knock signal voltage converting circuit 211 or the output voltage of the operational amplifier OP₁₁. In other words, it is so designed that the ignition timing is advanced a predetermined angle for every cycle while being retarded in accordance with the number of knock pulses.

Next, a description will be made of the operation of the output integrating circuit which controls the above-mentioned retard circuit 502, more particularly its starting action which provides a starting spark advance characteristic.

With the engine started through the operation of the starter, if the engine speed is below 350 rpm, the comparator CO₃ of FIG. 6 generates a high level output and the transistor T₄ is turned on by the output of the comparator CO₃ through the diode D₁₂ and the resistor R₂₉. When the transistor T₄ is turned on, the output (knock detection signal) from the comparator CO₁ of the comparator 208 shown in FIG. 3 is masked. As a result, the ignition timing is not retarded by the knock signal (including the noise signal). Also, the output of the comparator CO₁ is supplied to the output integrating circuit of the knock signal voltage converting circuit 211 of FIG. 7 through the resistor R₆₂ and the diode D₁₃ thereby selecting the starting maximum retard characteristic shown by the broken line in FIG. 10.

Next, the F-V generator 213 and the speed detecting circuit 214 shown in FIG. 6 will be described. The transistor T₆ of the F-V generator 213 is turned on in response to the establishment of two conditions that the output signal of the monostable circuit 212 goes to the high level and that the transistor T₉ is turned off. Thus, the transistor T₆ is turned on by the pulse of the width t₁ shown in (B) of FIG. 9. The period of this pulse is proportional to the engine speed and therefore the transistor T₆ is turned on and off in accordance with the engine speed. The voltage (about 1.7 V) at the junction point of the resistors R₆₇ and R₆₈ is applied to the positive terminal of the operational amplifier OP₁₄. When the transistor T₆ is turned on, a path including the capacitor C₁₉, the resistor R₈₄, the transistor T₆ and the ground is established from the output of the operational amplifier OP₁₄ and the capacitor C₁₉ is charged. When the transistor T₆ is turned off, the charge on the capacitor C₁₉ flows to the resistor R₈₅. The operational amplifier OP₁₄ generates an output corresponding to the difference between the voltages applied to its positive and negative input terminals and this output is applied to the negative input terminal of the comparators CO₃, CO₄ and CO₅, respectively. A fixed voltage (2.0 V) produced by the voltage division of the resistors R₃₃ and R₃₄ is applied to the positive input terminal of the comparator CO₃. Then, the voltage which is higher than 1.7 V and corresponding to the engine speed is applied to the negative input terminal of the comparator CO₃ and compared with the fixed voltage of 2 V. The output of the comparator CO₃ goes to the low level when the speed voltage is higher than 2 V and the output goes to the high level when the speed voltage is lower than 2 V. The voltage of 2 V forming a reference voltage corresponds to a low speed operation. More specifically, the engine speed corresponding to the voltage of 2 V is preset to 350 rpm. As a result, the output of the comparator CO₃ goes to the high level only when the engine speed is below 350 rpm.

On the other hand, a fixed voltage (3.0 V) produced by the voltage division of the resistors R₃₅ and R₃₆ is applied to the positive input terminal of the comparator CO₄. The voltage which is higher than 1.7 V and corresponding to the engine speed is applied to the negative input terminal of the comparator CO₄ and compared with the fixed voltage of 3 V. The output of the comparator CO₄ goes to the low level when the speed voltage is higher than 3 V and the output goes to the high level when the speed voltage is lower than 3 V. The reference voltage of 3 V corresponds to a high speed operation. More specifically, the engine speed corresponding to the voltage of 3 V is preset to 2,000 rpm. As a result, the output of the comparator CO₄ goes to the high level only when the engine speed is lower than 2,000 rpm. When the engine speed is below 2,000 rpm, the transistor T₈ of the knock signal voltage converting circuit 211 shown in FIG. 7 is turned on. When the transistor T₈ is turned on, the voltage applied to the negative input terminal of the operational amplifier OP₁₂ is decreased as compared with that applied when the transistor T₈ is off. Note that the purpose of the diode D₁₀ and the resistor R₄₆ is to provide a hysteresis characteristic and the reason for this is that since the knock signal voltage converting circuit 211 takes some time in responding to the speed of 2,000 rpm and the engine speed tends to increase somewhat during the interval, an output is generated which takes such speed increase into consideration.

On the other hand, the output of the operational amplifier OP₁₄ of the F-V generator 213 shown in FIG. 6 is applied to the negative input terminal of the comparator CO₅. A fixed voltage (5.0 V) produced by the voltage division of the series resistance of the resistors R₃₇ and R₃₈ is applied to the positive input terminal of the comparator CO₅. The voltage higher than 1.7 V and corresponding to the engine speed is applied to the negative input terminal of the comparator CO₅ and compared with the fixed voltage of 5 V. The output of the comparator CO₅ goes to the low level when the speed voltage is higher than 5 V and goes to the high level when the speed voltage is lower than 5 V. The reference voltage of 5 V corresponds to a high speed operation. More specifically, the engine speed corresponding to the voltage of 5 V is preset to 3,800 rpm. As a result, the output of the comparator CO₅ goes to the high level only when the engine speed is lower than 3,800 rpm. During the time that the output of the comparator CO₅ is at the high level, the base current of the transistor T₈ in the knock signal voltage converting circuit 211 of FIG. 7 is drawn by the transistor T₁₂. This is caused by the turning on of the transistor T₁₂ by the output of the comparator CO₅. Thus, when the engine speed exceeds 3,800 rpm, the output of the comparator CO₅ goes to the low level and the transistor T₁₂ is turned off. Consequently, the voltage supplied to the base of the transistor T₈ goes to the high level and the transistor T₈ is turned on. When the transistor T₈ is turned on, the voltage applied to the negative terminal of the operational amplifier OP₁₂ is decreased as compared with that applied when the transistor T₈ is off.

Next, the fail safe circuit 209 shown in FIG. 4 will be described. The fail safe circuit 209 detects an open or short condition and determines whether a background signal is present.

Firstly, the output of the half-wave rectifier circuit 205 is amplified by about ten times by the operational amplifier OP₇ and the resistors R₁₉ and R₂₀ In other words, if this amplification factor is represented by S₁, then the following is assumed ##EQU11##

The purpose of this amplification is to increase the resolution. In other words, when there is no knocking, a small signal is generated and this signal is amplified to detect a fault in the knock sensor. The amplified signal is compared with a fixed voltage (4.0 V) and it is generated as a pulse signal.

When there is an open fault in the knock sensor 100, no output is generated from the comparator CO₂. In the normal condition, the output of the comparator CO₂ goes to the high level in response to the output from the half-wave rectifier circuit 205. The capacitor C₉ is charged by the output of the comparator CO₂. As a result, the voltage at the junction point of the capacitor C₉ and the resistor R₈₉ is always high. In response to this high voltage, the transistor T₁₃ is turned on and the transistor T₁₄ is turned off. When the transistor T₁₄ is turned off, the resistor R₆₀ and the diode D₁₆ are not operated. Namely, nothing occurs when the knock sensor 100 is functioning normally.

When a fault occurs in the knock sensor 100, the comparator CO₂ generates no output (an open fault) or generates an output occasionally (a short fault). When the knock sensor 100 becomes faulty in this way, the capacitor C₉ is in a noncharged condition. As a result, the potential at the junction point between the capacitor C₉ and the resistor R₈₉ is always low. Therefore, the transistor T₁₃ shown in FIG. 7 is turned off so that in response to the turning off of the transistor T₁₃, the transistor T₁₄ is turned on and the transistor T₁₅ is turned off.

The transistor T₁₄ is also turned on and off in response to the output of the 2,000 rpm detecting circuit (the output of the diode D₁₁ shown in FIG. 6). In other words, when the engine speed is lower than 2,000 rpm, a high level output is generated from the diode D₁₁ and thus the fail safe circuit 209 comes into operation. When the engine speed is lower than 2,000 rpm and the terminal voltage of the capacitor C₉ is low, the transistor T₁₃ is turned off and the transistor T₁₄ is turned on. When this occurs, a current flow through the resistor R₆₀ and the diode D₁₆. As a result, the output of the operational amplifier OP₁₁ is caused by the capacitor C₁₈ to go to the high level. Simultaneously, the transistor T₈ is turned on and the ignition timing is retarded to the maximum retard angle.

In accordance with the present embodiment, there is thus no danger of the BGL being varied by any knock signal.

From the foregoing it will be seen that the present invention has the effect of improving the accuracy of knock detection. 

I claim:
 1. A knock control apparatus for internal combustion engines comprising:an amplifier for amplifying a signal generated from a knock sensor for detecting vibration of an engine; a filter for passing a knocking frequency range of an output from said amplifier; a rectifier circuit for rectifying an output from said filter; clamping means for clamping an output from said rectifier at a clamping voltage; background level detecting means for averaging an output from said clamping means; means for generating said clamping voltage at which said clamping means clamps the output from said rectifier on the basis of the average voltage from said background level detecting means so that said clamping voltage will continuously change in accordance with changes in the average voltage from said background level detecting means; and means for comparing the output from said filter with an output value from said background level detecting means to generate a signal for retarding an ignition timing in accordance with a knock intensity.
 2. An apparatus according to claim 1, wherein said filter is a band-pass filter.
 3. An apparatus according to claim 1, wherein said background level detecting means includes an operational amplifier whose output is applied to said clamping voltage generating means.
 4. An apparatus according to claim 1, wherein said background level detecting means includes an integration circuit having a resistor and a capacitor to average the output from said clamping means.
 5. An apparatus according to claim 1, wherein said clamping means includes a PNP type transistor having an emitter connected to an output circuit of said rectifier circuit, a base connected to an output circuit of said clamping voltage generating means and a collector connected to a predetermined reference potential.
 6. An apparatus according to claim 5, wherein said clamping voltage generating means includes a resistor and means for controlling an electric current flowing through said resistor on the basis of the averaged voltage from said background level detecting means, said base of the PNP transistor being supplied with a voltage occurring across the resistor.
 7. An apparatus according to claim 1, wherein said rectifier circuit is a half-wave rectifier. 